Wide band inductive coil device



Jan. 21, 1969 P. ALLEN Y WIDE BAND INDUCTIVE COIL DEVICE Sheet of 2 Filed Sept. 29, 1966 MEGAHERTZ INVENTOR. PHILLIP E. ALLEN ATTORNEY Jan. 21, T969 P. E. ALLEN 3,423,710

WIDE BAND INDUCTIVE COIL DEVICE I Filed Sept. 29, 1966 Sheet g of 2 e44 f Posm'vE 57 R39 SWEEP GENERATOR R053! SIGNAL 38 R40 GENERATOR POSITIVE PULSE SWEEP SOURCE GENERATOR INVENTOR.

PHILLIP EALLEN ATTORNEY:

United States Patent 3,423,710 WIDE BAND INDUCTIVE COIL DEVICE Phillip E. Allen, Reno, Nev., assignor t0 the United States of America as represented by the United States Atomic Energy Commission Continuation-impart of application Ser. No. 429,934,

Feb. 2, 1965. This application Sept. 29, 1966, Ser. No. 593,254 US. Cl. 336-155 6 Claims Int. Cl. Htllf 21 /08 ABSTRACT OF THE DISCLOSURE This application is a continuation-in-part of application Ser. No. 429,934, filed Feb. 2, 1965, now abandoned.

The present invention was evolved in the course of, or under, Contract No. W-7405-ENG-48 with the United States Atomic Energy Commission.

The present invention relates, in general, to wide band inductive coil devices utilized in electrical circuitry, and more particularly to an inductive coil device wherein ferromagnetc core elements of preselected core loss and permeability characteristics are arranged to cooperate with solenoidal windings to provide a compensated, uniform, wide megacycle band frequency response.

Inductive coil devices utilizing single coil or multiple mutual inductance-linked coils, including inductors and particularly transformers, are operated at megacycle frequencies, e.g., in the range of 1 to 200 megacycles, for instrumentation, communication and other purposes. Devices available heretofore have imposed drastic limitations on the range of frequencies which could be passed simultaneously therethrough with uniform response characteristics in a single channel device. In the earlier days of high frequency equipment design, air core solenoids were generally employed, but such devices are inherently limited in bandwidth, effective inductance and the like, to the extent that only exceedingly complex low efficiency circuits could give bandwidths of even minimal effectiveness in the megacycle ranges.

With the advent of powdered iron, and then later with the advent of the ferrite-type ferromagnetic low loss core materials, notable improvements were achieved during the recent past in the performance of inductive core devices used at high frequencies. Notwithstanding these advances in magnetic core materials, inductive electrical coil devices, including transformers, inherently have had an inordinately narrowly defined and limited frequency range of useful operation. The limited frequency range of such devices unduly restricts the useful operating frequency range of associated equipment, which equipment could otherwise have a much broader application and could be of greatly simplified and more versatile design. In order to change the frequency range of operation of equipment, it has been the practice to manually change coils. Alternately, sequential switching system arrangements or elaborate crossover circuitry employing a plurality of coils, each having a different useful frequency range, may be resorted to in efforts to solve the problem, however, at the expense of creating other problems such as cross-talk or reliability problems generally encountered with the use of complex circuitry.

While such expedients may be utilized in single chan- 3,423,710 Patented Jan. 21, 1969 ice nel operation where the phenomena observed are of a narrow bandwidth within the individual bandwidth capability of the switched channels or the communication can be repeated, it will be observed that a plurality of overlapping channels connected in parallel relation is required for simultaneous reliable treatment of all portions of a wide bandwidth signal, or of a signal which may occur at any unpredetermined portion over a wide range of frequencies, e.g., 1 to 200 megacycles as produced by a nonreproducible source such as signals produced by a natural phenomenon, instrumented explosions, etc.

It has now been discovered that, by appropriate selection of core materials and arrangements of the core materials to form a composite core, there results a cooperative interaction between the core and winding or windings of such an inductive coil device so as to provide a greatly extended uniform frequency response bandwidth. Briefly stated, the inductive coil device of the invention essentially includes a plurality of annular magnetic cores of selected magnetic permeabilities and resistive loss characteristics which are disposed immediately adjacent to each other in coaxial alignment. Each winding, one or more of which may be utilized as in conventional practice to provide an inductor, autotransformer, transformer or the like, are disposed so that each and every turn of each winding encircles the composite core in coextensive parallel relation to provide mutual induction effective for cooperative action between said bores and windings. At least two core elements having distinct permeabilities and core loss characteristics are required to give comparatively constant relative impedance and uniform response. As a general criterion it may be said that one of the cores is selected by characteristics appropriate to the lower portion of the desired band coverage, and the second as appropriate to the upper portion of the desired band coverage. With very wide bandwidth, due to inherent limitations of each, one might expect that a gap in frequency response between such portions would exist. However, quite surprisingly, it is found that the arrangement provides uniform response extending continuously over the frequency ranges of each individual core section, as well as the frequency range between them. Thus, by combining cores of preselected permeabilities and core loss functions and appropriate windings, I have been able to provide a transformer, e.g., which operates over a frequency range between at least 5 and 200 megacycles per second. Accordingly, by virtue of the proper selection of permeability and core loss characteristics, the frequency range of inductive coil devices constructed in accord with the foregoing teachings far surpasses the useful range of devices previously available. The present coil structures can therefore be employed in numerous applications without resort to coil exchanging and switching techniques mentioned above, and in the use of a single channel wide band instrument instead of a device utilizing multichannel systems. Appropriate application of the invention should be apparent to those skilled in the art. However, information as to details of some such applications are described in my paper entitled Multi-Core Transformers Boost Bandwidth, P. Allen, Electronic Design, Feb. 3, 1964.

Composite core structures using materials of differing permeabilities for constructing inductive devices have been contemplated in the prior art for other purposes. For example, U.S. Patent No. 1,849,845, issued to C. G. Maya on Mar. 15, 1932, discloses an instrument or current transformer, evidently for use with conventional AC power systems, in which two laminated core sections punched from sheet metal are disposed in spaced coaxial relation with an auxiliary winding, in addition to conventional primary and secondary windings, connected in opposing series, i.e., bucking relation so that a core permeability intermediate from the two elements forming the composite is provided. Another composite core in which a different permeability, using core elements of differing permeabilities, is disclosed by Ehlers in US.

core was designed to provide Pupm, i.e., loading inductor frequency AC systems, and no recognition of the possibility of, or the teachings of, the provision of a wide plateau uniform response at megacycle frequencies appears therein.

Generally stated, it is a major object of this invention to provide a coil having a wide frequency range of useful operation.

'A further object of the invention is to provide a transformer having a uniform response over a wide range of frequencies.

Another object is to provide a wide band transformer which is relatively simple to fabricate and is inexpensive.

A further object of the present invention is to provide a transformer including a plurality of cores constructed of materials having different selected permeabilities and core loss characteristics interacting with the winding thereon to provide an extended uniform frequency bandwidth.

These and other objects of the present invention will become more apparent in light of the following description and the accompanying drawings, in which:

FIGURE 1 is a perspective view of a wide band transformer inductive coil device of the preferred embodiment of the present invention;

FIGURE 2 schematically depicts an equivalent circuit of the transformer of FIGURE 1 which is useful for an analysis of the electrical and magnetic properties of the device;

FIGURE 3 graphically depicts the electrical response of the transformer of FIGURE 1 to an imposed megacycle range alternating current electrical signal; and

FIGURE 4 is a circuit diagram of a Rossi sweep system incorporating the wide band transformer.

A discussion of the factors well understood in the art which determines the useful operating frequency band of an inductive coil device will provide background for understanding of the invention. First, it is a characteristic of ferromagnetic cores used in such inductive coil devices that core losses increase as a function of the frequency of a current traversing the coil winding, thereby defining an approximate high frequency limit, or cutoff point, above which the coil becomes ineffectual as an inductive element used as an inductor, transformer primary or the like. In addition, it is typical that, of core elements which are currently available, the cores of higher permeability exhibit higher core losses than lower permeability ferromagnetic cores for the same frequency. Secondly, the inductive impedance of the windings, which is a direct function of both core permeability and frequency, must be large enough at low frequencies to develop a suflicient winding voltage. This becomes particularly important, for example, when a transformer is employed in circuits where a signal is to be communicated from a primary winding to a secondary winding of the transformer, or to couple stages of a circuit. Now, if a core material of low enough permeability to operate Without excessive core losses at higher frequencies is used, as in conventional practice, the primary winding inductive impedance be- :omes too small at lower frequencies to develop the full winding voltage and thus the input signal is greatly at- :enuated. Conversely, if a core material having a high :nough permeability to provide sufficient winding inluctance at the lower frequency is used, then its core Osses are excessive when operated at the high frequen- Car cies. Accordingly, each core material has a limited useful frequency band width defined in general by its permeability and core loss characteristics. Thus, in accordance with conventional design practice, it is generally recognized that the use of either material alone in a wide band transformer provides only a limited frequency band of uniform response.

However, as now discovered, by combining core elements of the types discussed above which provide effective or optimum operation at widely separated frequencies into a composite core in such an inductive coil structure, the elements of the composite core and windings interact to provide overall frequency response bandwidths which extend beyond the normal low frequency limit of the low frequency core, and above the high frequency limit of the high frequency core bandwidths of the individual cores used by themselves. In order to extend the bandwidth of uniform response of the inductive coil device using such a composite core, the frequency characteristic curves of the individual cores are selected and arranged to overlap in a predetermined fashion in the region intermediate between the high and low frequency ranges of the individual cores. The frequency range and shape of the frequency response characteristic curve is a function of both permeability and core loss characteristics. The desired characteristics are obtained by selecting individual cores having permeabilities and core loss characteristics appropriate to provide individual frequency response curves in the high and low frequency portions of the frequency range desired and then combining these cores into a single composite core structure with coextensive winding provided in a series magnetic circuit relationship essential for proper operation. This relationship will be more fully described below.

In brief, the construction of a transformer according to the invention includes at least two closed loop cores individually selected according to convention design criteria to have different useful frequencies of operation, as described below. These cores are then placed side-byside in coaxial alignment to form a single composite core. Then windings, e.g., a primary and a secondary, are disposed on this composite core with each and every turn of said windings being unidirectional and coextensively encircling the loop core. The product is a transresults in a relatively flat frequency response over wide selected frequency ranges. For example, it is often desirable to provide coils or transformers, wherein the frequency response over an extensive range does not deviate from the mean response more than 3 decibels. Note More particularly, in FIGURE 1, there is illustrated a wide band transformer including a composite ferromagnetic core 11 comprising a first annular loop core 12 and a second annular loop core 13, disposed contiguously and coaxially aligned, one having a low operating frequency range and the other relatively high operating frequency range defining generally the lower and upper range limits of the desired bandwidth. A primary winding 14 encircles both members of composite core 11, as does at least one secondary winding 16. As indicated above, each winding is unidirectional and each turn of each winding encircles the loop core coextensively with each other turn passing through the open center of the core and generally normal therearound. Winding 16 is provided with appropriate terminals 17 and 18 to connect to an electrical load 19. Primary winding 14 is provided with a pair of terminals 21 and 22 to connect to an appropriate alternating current signal source 23.

In one embodiment, a Ferroxcube No. 102 toroid which had an initial permeability of approximately 250 and core loss characteristics defining a maximum frequency limit of about 12 megacycles was selected for core 12, while a Ferroxcube No. 1Z2 toroid with an initial permeability of approximately and core loss characteristics defining a maximum frequency limit of about 200 megacycles was selected for core 13, both of these cores being manufactured and distributed by Ferroxcube Corporation of America, Saugerties, N.Y. Two turns of No. 25 enameled copper wire was used for primary winding 14, while 3 turns of No. 25 enameled copper wire was used for secondary winding 16. It is noted that the number of turns used is relatively low to avoid excessive capacitance leakage between the windings in the megacycle frequency range. The inner and outer diameters of both toroids were approximately 3 mm. and 8 mm., respectively, while the thickness of the No. 102 and No. 1Z2 toroids were approximately 6 mm. and 5 mm., respectively. With this arrangement, a 3-decibel frequency bandwidth from 5 megacycles per second to 200 megacycles per second was obtained. A comparable transformer using only the Ferroxcube No. 102 core had a limited usable frequency range from 5 megacycles per second to 12 megacycles per second. Similarly, a transformer using only the Ferroxcube No. 1Z2 core has a limited usable frequency range from 50 megacycles per second to 200 megacycles per second. It is significant that the inductive coil device of this invention provides an effective response characteristic not only over the frequency bands of the individual cores, but also, most remarkably, provides an effective uniform response in the intermediate frequency region not covered by either of the cores when used individually. Magnetic core materials suitable for use in the present invention are those which are generally usable over the range of low megacycle, i.e., above about 1 megacycle to at least 100 megacycles and above. A considerable range of materials are accordingly available for selecting the low and high frequency cores according to the teaching of the invention. Preferred materials are the ferrites such as Ferroxcube, Ceramag and Ferromic which are mixtures of manganese, nickel and zinc ferrites and are manufactured and sold in a variety of shapes and grades. The characteristics of typical ferrites are disclosed, e.g., at pages 217218, Electronic Transformer and Circuits, Second Edition, John Wiley and Sons. At the lower frequencies, bonded powdered iron cores might possibly be used. In any event, ferrites having permeabilities of the order of 10-5000 and internal resistances of the order of 10 to 10 ohms per cubic centimeter aresuitable, as would be other equivalent ferromagnetic materials and especially those adapted for high radio frequencies. Cores having higher permeabilities and lower resistances in said ranges are generally suitable as low frequency cores, and those with low permeabilities, e.g., 10 to 20, and high resistances are suitable as high frequency cores.

When two cores are disposed and with windings provided as shown in the preferred embodiment of FIGURE 1, the primary winding impedances of the cores are effectively in electrical series with each other, although the cores are physically disposed in seeming parallel relationship. For the purposes of analysis, the simplified equivalent circuit of FIGURE 2 represents the two component core transformer shown in FIGURE 1. This circuit shows the equivalent circuit of the transformer as seen from the primary winding 14. Inductors 24 and 26 having respective inductances L and L represent equivalent inductance values for cores 12 and 13 respectively. Resistors 27 and 28 having respective resistance values R and R represent core losses for these respective cores 12 and 13. Transformer 29 represents an ideal transformer which has an infinite impedance for the purpose of this analysis. Core 12 is selected with a permeability p12 to provide substantial impedance, characterized by inductor 24, at low frequencies in a selected wide band frequency range, together with core losses, characterized by resistor 27, which effectively short out the impedance provided by inductor 24 at intermediate frequencies within the selected range. Core 13 is selected with a permeability 1. of value less than to provide substantial impedance, characterized by inductor 26, at higher frequencies in the selected range, which, together with core losses characterized by resistor 28, provide the maximum high frequency limit for the selected range. It is noted that the other parameters affecting coil impedance, namely winding turns and core dimensions, are generally confined to a narrow range of values because of practical consideration. Therefore, the core permeabilities p12 and #13 are treated above as the sole variables in fixing the values of inductors 24 and 26.

At this point, it may be well to set forth some well known transformer design formulas which are pertinent to the present invention. First the approximate inductance L of a toroid with an outer diameter d an inner diameter d and a thickness h is given by the following equation which also appears in the book by H. H. Skilling, entitled Electrical Engineering Circuits, John Wiley & Sons, Inc., 1960.

where N is the number of turns of the particular winding on the core and where p. is the permeability of the core material. Secondly, the bandwidth characteristics of coils is expressed in terms of a quality factor, Q, which is defined by the ratio of the equivalent parallel core less resistance, R (R being analogous to R or R of FIG- URE 2), to the inductance reactance, wL, i.e.,

where w:21r times the frequency of the signal imposed on the transformer winding. Q and R of commercially available cores may be determined from manufacturers design specifications (General Information on Ferrites, Bulletin 101; Ferroxcube Corporation of America, Saugerties, N.Y.). These parameters may also be determined empirically by conventional bridge and null methods. Note that in characterizing the bandwidth of coils, the quantity factor Q is a function of the variables upon which core permeability and core losses depend, i.e., wL and R respectively. Hence the choice of magnetic cores, normally thought of in terms of magnetic permeability and losses, may be specified in terms of quality factor Q. Another method of characterizing the bandwidth of coils is in terms of input impedance. This is explained by the fact that the same variables which specify the quality factor Q, i.e., wL and R also specify input impedance. FIGURE 3 shows plots of wL and R for two cores and a dashed line representing the vector summation of wL and R Thirdly, it may be helpful to remember that the ratio of the output voltage of a transformer to its input voltage is very nearly proportional to the ratio of the number of turns on the secondary Winding to the number of turns on the primary winding. For the purpose of this writing, the usable frequency band of a transformer may be defined roughly as those frequencies over which the actual output voltage of the transformer does not deviate to an intolerable degree from that expected output voltage as calculated by multiplying the input voltage by the above-mentioned turns ratio. Also, for purposes of this writing, the term transformer response may be taken to mean the ratio of the actual transformer output voltage to the expected calculated output voltage. As will be shown below, this response ratio varies, and can be plotted as a function of frequency.

As a further point of clarification, it may be noted that this discussion adopts the standard definition of the term ferromagnetic. This ferromagnetic refers to those materials whose relative permeabilities significantly exceed unity and are usually at least several times unity. The term permeability, as used herein, relates to either absolute permeability or specific permeability as computed from the ratio of magnetic flux induction to magnetizing force, the computation being made, of course, within a consistent set of dimensional units.

FIGURE 3 graphically portrays the impedance variation as a function of frequency of each of the elements shown in FIGURE 2. Since #12 has been selected to be substantially larger than then wL is greater than (01. for cores having approximately equal dimensions, where w is defined as above. The core loss, which is represented by resistors 27 and 28, is quite nearly inversely proportional to the frequency.

Analysis of FIGURES 2 and 3 shows that at the low frequencies, inductor 24 alone develops the primary winding voltage. As the frequency rises, resistance R decreases and would cause considerable core loss but for the fact that the impedance wL increases to prevent this loss. As the frequency becomes even higher, R decreases and hence bypasses inductor 26, allowing losses in both resistor 27 and resistor 28. Hence, it is seen that inductor 24 limits the lowest useful frequency, and resistor 28 in conjunction with stray capacitances not heretofore mentioned determines the highest usable frequency. Resistors 27 and 28 and inductors 24 and 26, as electrically combined, coact to produce the equivalent impedance Z of the transformer. The dotted line, flagged as Z in FIGURE 3 shows this equivalent impedance of the transformer as a function of frequency. The frequency response of the transformer, excluding external effects, is roughly proportional to this impedance Z with a driving source impedance near infinity. Of course, if a lesser driving point impedance is used, the frequency response will be further enhanced. Also, it is noted that L and L may be varied by changing the dimensions of cores 12 and 13 and/or the number of primary winding turns, within practical limits, to achieve a desired shape of the wide band impedance response characteristics.

In FIGURE 3, the curve Z is the vector summation of resistances and inductive reactances of the two cores at individual frequencies from O to 400 mH Z reaches localized maxima (designated Z and Z near the opposite bounds of Z i.e., and 100 mH (designated f and f Note that between Z and Z the impedance hereafter designated Z does not significantly vary.

To illustrate the exceptional performance of such a wide band transformer, it has been advantageously used in a Rossi sweep system. In general, a Rossi system entails superimposing a single, fixed frequency sine wave timing signal on an oscilloscope pulse trace to provide accurate measurement of the duration, amplitude and steepness of the trace. To permit satisfactory analysis of a wide range of pulse shapes, a plurality of tuning signals are used wherein each of these signals is supplied by a separate crystal-controlled oscillator called a Rossi oscillator. The frequencies of the timing signals have been standardized at 5, 10, 20, 40, 50, 80, 100 and 200 megacycles.

Now referring to FIGURE 4, a Rossi signal generator 30 comprised of a plurality of individual Rossi oscillators is connected across a serial arrangement of compensating network 31, variable resistor 32 and a primary 33 of wide band transformer 34. Compensating network 31 is comprised of a power limiting resistor 35 in parallel with high frequency bypass capacitor 36. Transformer 34, secondaries 37 and 38, each having a first and second end, are provided with load resistors 39 and 40, respectively. An LC series network 41 comprising inductor 42 and capacitor 43 is connected between the first ends of said secondaries 37 and 38. Capacitors 44 and 45 are individually connected between the first ends of secondaries 37 and 38 and ground. The second ends of the secondaries 37 and 38 are attached to opposing deflection plates of oscilloscope 46. The pulse source 47 to be analyzed is connected acres the remaining pair of deflection plates as shown. Positive sweep generator 48 is electrically communicative with the first end of secondary 37 through resistor 49, while negative sweep generator 50 is connected to the first end of secondary 38 by means of resistor 51.

In the operation of the circuit of FIGURE 4, Rossi signal generator 30 is set to provide one of the standard Rossi frequency signals (i.e., 5, 10, 20, 40, 50, 80, or 200 mc./sec.). The Rossi signal is mixed with a positive or negative sweep signal emanating from positive sweep generator 48 or negative sweep generator 40 by means of transformer 34. Therefore, one pair of deflection plates of oscilloscope 46 receives a sweep signal having a Rossi Sinusoid wave superimposed thereon, while the remaining deflection plates are responsive to the pulse source 47.

Resistor R35 is necessary in applicants circuit to protect the transformer 34 from the maximum power output of the Rossi generator 30. Without R35, transformer 34 may overheat causing degradation in the permeability characteristics thereof. Bypass capacitor 36 shorts out R35 at the higher frequencies, permitting a larger voltage drop across primary 33, thereby extending the frequency range of the circuit. Variable resistor R32 prOvides for variation in the amplitude of the output signal. Resistors R39 and R40 serve as dissipative loads for the secondary windings 37 and 38. Capacitors C44 and C45 are used to provide an AC ground for the secondary windings and also to form a pi filter with resistors R49 and R51 and the output capacitance of the sweep generators 48 and 50. The purpose of the pi filter is to isolate the Rossi signal from the sweep generators 48 and 50.

Series LC network 41 is adjusted to provide a low impedance path between the first ends of secondaries 37 and 38 at 200 mc./sec. This enables the secondaries 37 and 38 to drive the output capacitors C44 and C45 which exhibit only slight impedance at 200 mc./ sec.

The following component Values should be used to reproduce the above circuit:

Rossi oscillator Output voltage 42 volts. R35 100-400 ohms. C36 4.7 pf. R32 100K ohms. R39 and R40 1K ohm. R49 and R51 100 ohms. C44 and C45 l0 pf. C42 .2 117. C43 1.27 pf. Transformer 34:

Cores Ferroxcube No. 102thickness 6 mm., No. 1Z2- thickness 5 mm. Turns One primary, 2 turns No. 25 enameled copper wire,

two secondaries, 3 turns No. 25 enameled copper wire.

As described herein, the transformer cores are selected and combined to give an approximately flat response over a band of frequencies. However, it is possible to combine a larger plurality of cores of selected materials to produce a single transformer having discreet pass bands with other selected frequency-dependent transmission characteristics. Also, it should be noted that the composite core described herein need not be limited in its use to a transformer having a primary and secondary winding. Such a composite core can be used as a core for a wide band choke coil or auto-transformer. Although the invention has been described above in terms of a preferred embodiment, the invention should be construed liberally, and it will be understood that various changes and modifications may be made without department from the spirit and scope of the invention.

What is claimed is: 1. An inductive coil device having an effective impedance over a wide band frequency range, comprising:

(a) a first closed loop core of ferromagnetic material having a high permeability in the range 200400;

(b) a second closed loop core of ferromagnetic material in proximate coaxial alignment with said first core, said second core having a lower permeability in the range -30; and

(c) at least a first unidirectional electrical conductor winding encircling both said cores and interacting therewith, wherein said first ferromagnetic core interacting with said conductor winding, is selected to have a first input impedance Z whose maximum value is attained at a first frequency f said second toroidal magnetic core interacting with said conductor winding is selected to have a second input impedance Z whose maximum value is attained at a second frequency f where Z approximates Z but where h and f are near the opposite extremities of said wide bandwidth, and said magnetic cores interacting with said conductor further selected to jointly have a combined impedance Z approximating Z and Z in those frequency regions between f and f 2. A device as defined in claim 1, wherein a second electrical conductor winding unidirectionally encircles both said cores, thereby both said cores interact with both said windings, presenting a wide band transformer where said first winding provides a primary and said second winding provides a secondary for said transformer.

3. An inductive coil device as defined in claim 1, Wherein each of said cores defines a toroidal configuration, and said cores are coaxially juxtaposed.

4. An inductive coil device as defined in claim 1, wherein:

(a) said first closed loop core has in conjunction with said winding a first impedance frequency response bandwidth over which the impedance Z provided individually by said first core and said winding remains within three decibels of a mean impedance response thereof;

(b) said second closed loop core has in conjunction with said winding a second impedance frequency response bandwidth over which the impedance Z provided individually by said second core in said Winding remains within 3 decibels of a mean impedance response thereof.

5. A device as defined in claim 4, wherein said first winding provides a primary of a transformer and at least a second electrical conductor winding is unidirectionally wound around both said cores providing a secondary winding for said transformer whereby said interaction provides a wide band frequency response transformer.

6. A device as defined in claim 4, wherein each of said cores define cross sectionally segmented hollow cylinders having equal radial dimensions, and said cores are coaxially contiguously juxtaposed.

References Cited UNITED STATES PATENTS 1,748,857 2/1930 Wellings et al. 336 XR 1,812,740 6/1931 Ehlens 336-2l2 1,849,845 3/1932 Mayo 336-182 1,971,207 7/1934 Boyajian et al. 336-l74 XR LEWIS H. MYERS, Primary Examiner.

T. I. KOZMA, Assistant Examiner.

US. Cl. X.R. 336--212 

